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tm 1 file number 4633.2 HFA3783 i/q modulator/demodulator and synthesizer the HFA3783 is a highly integrated and fully differential sige baseband converter for half duplex wireless applications. it features all the necessary blocks for quadrature modulation and demodulation of ??and ??baseband signals. it has an integrated agc receive if ampli?r with frequency response to 600mhz. the agc has 70db of voltage gain and better than 70db of gain control range. the transmit output also features gain control with 70db of range. the receive and transmit if paths can share a common differential matching network to reduce the ?ter component count required for single if half duplex transceivers. a pair of 2nd order antialiasing ?ters with an integrated dc offset cancellation architecture is included in the receive chain for baseband operation down to dc. in addition, an if level detector is included in the agc chain for threshold comparison. up and down conversion are performed by doubly balanced mixers for ??and ??if processing. these converters are driven by a broadband quadrature lo generator with frequency of operation phase locked by an internal 3 wire interface synthesizer and pll. the device operates at low lo levels from an external vco with a pll reference signal up to 50mhz. the HFA3783 is housed in a thin 48 lead lqfp package well suited for pcmcia board applications. features integrates all if transmit and receive functions broad quadrature frequency range . . . . . .70 to 600mhz 600mhz agc if strip with level detector . . . . . . . . .69db dc coupled baseband interfaces integrates a receiver dc offset calibration loop integrated 3 wire interface pll for lo applications low lo drive level . . . . . . . . . . . . . . . . . . . . . . . -15dbm fast transmit-receive switching . . . . . . . . . . . . . . . . <1 s power management/standby mode single supply 2.7 to 3.3v operation applications ieee802.11 1 and 2mbps standard systems targeting ieee 802.11 11mbps standard wireless local area networks pcmcia wireless transceivers ism systems tdma packet protocol radios simpli?d block diagram ordering information part number temp. range ( o c) package pkg. no. HFA3783in -40 to 85 48 ld lqfp q48.7x7a HFA3783in96 -40 to 85 tape and reel 0 o /90 o pll module ref in if 2x lo / vco in 3 wire interface baseband tx i baseband txq transmit if agc baseband rxi baseband rxq receive agc if detector out charge pump out if_in if_out offset cal cal enable i q data sheet march 2000 caution: these devices are sensitive to electrostatic discharge; follow proper ic handling procedures. 1-888-intersil or 321-724-7143 | intersil and design is a trademark of intersil corporation. | copyright intersil corporation 2000 prism is a registered trademark of intersil corporation. prism logo is a trademark of intersil corporation.
2 pinout pin descriptions pin number name description 1 rx_v cc receive agc ampli?r power supply. requires high quality capacitor decoupling. 3 if_rx+ receive agc differential ampli?r non-inverting if input. requires a dc blocking capacitor. 4 if_rx- receive agc differential ampli?r inverting if input. requires a dc blocking capacitor. pins 3 and 4 are interchangeable and can be used single ended with the other being capacitively bypassed to ground. 6 tx_vagc transmit agc ampli?r dc gain control input. 7 tx_v cc transmit agc ampli?r power supply. requires high quality capacitor decoupling. 8 if_tx+ transmit agc differential ampli?r positive output. open collector requiring dc bias from v cc through an inductor. 9 if_tx- transmit agc differential ampli?r negative output. open collector requiring dc bias from v cc through an inductor. 10 tx_v cc transmit agc ampli?r power supply. requires high quality capacitor decoupling. 13 ref_byp pll reference buffer signal negative differential input. pin has active bias and can be used in conjunction with pin 14 either differential or single ended. cmos inputs must be dc coupled. small sinusoidal inputs must be dc blocked with this pin bypassed to ground via a capacitor. 14 ref_in pll reference buffer signal positive differential input. pin has active bias and can be used in conjunction with pin 13 either differential or single ended. cmos inputs must be dc coupled. small sinusoidal inputs must be dc blocked with this pin used as an input for the reference signal. when used with single ended cmos inputs, pin 13 must be left ?ating. pins 13 and 14 are interchangeable. 17 syn_v dd pll synthesizer digital power supply. requires high quality capacitor decoupling. 18 clk pll synthesizer serial interface clock. cmos input. 19 data pll synthesizer serial interface data. cmos input. 20 le pll synthesizer serial interface latch enable control. cmos input. 1 2 3 4 5 6 7 8 32 31 30 29 28 27 26 25 24 23 22 21 20 19 18 17 9 10 11 12 13 14 15 16 33 34 35 36 37 38 39 40 41 42 43 44 45 46 47 48 rxq+ rxq- txi+ txi- 1.2v_out txq+ txq- gnd lo_v cc lo_in+ lo_in- gnd rx_v cc gnd if_rx+ ir_rx- gnd if_tx+ if_tx- tx_v cc gnd gnd tx_vagc tx_v cc gnd rx_vagc gnd if_det pe1 cal_en gnd bb_v cc gnd rxi+ rxi- pe2 ref_byp ref_in gnd gnd syn_v dd clk data le cp_v dd cp_d0 gnd ld HFA3783
3 21 cp_v dd pll charge pump power supply. independent supply for the charge pump, not to exceed 3.6v. requires high quality capacitor decoupling. 22 cp_d0 pll charge pump current output. 24 ld pll lock detect output. requires low capacitive loading not to exceed 5pf. 26 lo_in- local oscillator differential buffer negative input. requires ac coupling. for single ended applications its complementary input, pin 27, must be bypassed to ground via a capacitor. 27 lo_in+ local oscillator differential buffer positive input. requires ac coupling. for single ended applications its complementary input, pin 26, must be bypassed to ground via a capacitor. pins 26 and 27 are interchangeable. note: high second harmonic content lo waveforms may degrade i/q phase accuracy. 28 lo_v cc local oscillator buffer ampli?r power supply. requires high quality capacitor decoupling. 30 txq- baseband quadrature differential inputs for if transmission. dc coupled requiring 1.3v common mode bias voltages. 31 txq+ 32 1.2v_out highly regulated band gap 1.2v buffered output. used in conjunction with adcs and dacs for voltage /temperature tracking. requires high quality 0.1 f capacitor decoupling to ground. 33 txi- baseband in phase differential inputs for if transmission. dc coupled requiring 1.3v common mode bias voltages. 34 txi+ 35 rxq- baseband quadrature differential outputs from if demodulation. dc coupled output with 1.2v common mode dc outputs. ac coupling pins 35, 36, 37 and 38 requires programmable register activation for dc hold during tx to rx switching. 36 rxq+ 37 rxi- baseband in phase differential outputs from if demodulation. dc coupled output with 1.2v common mode dc outputs. 38 rxi+ 40 bb_v cc baseband receive lpf output and offset control power supply. requires high quality capacitor decoupling. 42 cal_en cmos input for activation of internal dc offset adjust circuit for the receive baseband outputs. a rising edge activates the calibration cycle, which completes within a programmable time and holds the calibration while this pin is held high. in applications where the synthesizer is not used, this pin needs to be grounded. 43 pe2 power enable control pins: please refer to the power enable truth table in the electrical speci?ations section. 44 pe1 45 if_det if detector current output. a current source of 175 a typical is generated at this pin when the if agc receive differential or single ended signal at pins 3 and 4 is between 100 and 200mv pp . 47 rx_vagc receive agc ampli?r dc gain control input. 2, 5, 11, 12, 15, 16, 23, 25, 29, 39, 41, 46, 48 gnd grounds. connect to a solid ground plane. pin descriptions (continued) pin number name description HFA3783
4 application circuit 100p 21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 48 47 46 45 37 38 39 40 44 43 42 41 17 18 19 20 0/90 synth 2k 100p 100p 1000p 1000p 49.9 1000p 0.01 0.1 0.022 0.22 2k 3.92k 100p 0.1 0.1 0.01 100p 2.87k l p l p c s c s vt 0.1 56 0.01 56p 124 124 536 124 124 536 619 976 idac 7 bits tx_vagc idac 7 bits rx_vagc 1-bit if_det adc 6 bits rx? adc 6 bits rx? dac 6 bits tx? tx? 1.2v ref in hfa3861 from mac (cal+ en ctrl) vco_v cc vco panasonic enfv25f80 rf from mac (pll ctrl) ref freq (sinusoidal) 10 0.1 10 v cc lo saw det dac 6 bits 1000p 3900pf 68p 68p sawtek 855653l1 HFA3783
5 test diagram 21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 48 47 46 45 37 38 39 40 44 43 42 41 17 18 19 20 100p 100p 270p 1000p 1000p .01 .1 100p 100p .1 .1 .01 2.87k .1 50 .01 270p 56p (sinusoidal) 10 v cc lo common mode voltage calibration calibration 1.2v ref. calibration calibration tc4-1w 2k 8p 8p 27n 27n 1000p match components for if in/out ref_in 50 clk data le vcc/2 cp (low input capacitance) lo_in (2x freq) 1.2v_out rx_vagc if_det pe1 pe2 cal_en txq v cc 50 50 1000p 1000p frequency response test set up 50 200p 50 analyzer sweep 9 8 7 5 4 3 2 6 gen. 0/90 synth 5k ? input rxi 5k ? input rxq common mode voltage txi buffer test fixture (374mhz) and transformer tx_vagc 100p 1000p HFA3783
6 absolute maximum ratings thermal information voltage on any other pin. . . . . . . . . . . . . . . . . . . -0.3 to v cc +0.3v v cc to v cc decouple or gnd to gnd . . . . . . . . . . . . . -0.3 to +0.3v any pin to gnd. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.0v operating conditions operating temperature range . . . . . . . . . . . . . . . . . . -40 to +85 o c supply voltage range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.7-3.3v thermal resistance (typical, note 1) ja ( o c/w) jc ( o c/w) lqfp package . . . . . . . . . . . . . . . . . . . 70 n/a maximum junction temperature (plastic package) . . . . . . . . . . 150 maximum storage temperature range . . . . . . . . . . . . . . . -65 to 150 maximum lead temperature (soldering 10s) . . . . . . . . . . . . . . . 300 moisture sensitivity level (intersil tech. brief tb363). . . . . .168 hrs caution: stresses above those listed in ?bsolute maximum ratings may cause permanent damage to the device. this is a stress only rating and operatio n of the device at these or any other conditions above those indicated in the operational sections of this speci?ation is not implied. note: 1. ja is measured with the component mounted on an evaluation pc board in free air. dc electrical speci?ations parameter temp. ( o c) min typ max units supply voltage full 2.7 - 3.3 v receive total supply current 25 - 36 40 ma transmit total supply current 25 - 32 40 ma voltage reference output at 1ma, 0.1 f load full 1.14 1.2 1.26 v note: tx/rx power down supply current (pll serial interf. active) (note 2) full - - 100 a tx/rx/power down speed (note 3) full - - 10 s rx/tx, tx/rx switching speed (note 3) full - - 1 s cmos low level input voltage full -0.3 - 0.3*v dd v cmos high level input voltage (v dd = 3.6v) full 0.7*v dd - 3.6 v cmos threshold voltage full - 0.5*v dd -v cmos high or low level input current full -3.0 - +3.0 a note: 2. standby current is measured after a long elapsed time (20 seconds). 3. tx/rx/tx switching speed and power down/up speed are dependent on external components. receive cascaded ac electrical speci?ation if = 375mhz, lo = 748mhz, v cc = 2.7v, unless otherwise speci?d parameter test conditions temp. ( o c) min typ max units if frequency range test diagram full 70 - 600 mhz 2xlo frequency range test diagram full 140 - 1200 mhz maximum power gain vagc = 0v 25 56 61 db voltage gain nominal high gain. differential 250 ? in, 5k ? output differential load. agc control voltage set to 69db of voltage gain full - 69 - db power gain full - 56 - db cascaded noise figure full - - 8 db output ip3 full +2.2 - - dbm output p1db full -14.1 - - dbm HFA3783
7 voltage gain agc control voltage set to 10db attenuation. differential 250 ? input, differential 5k ? output load. full - 59 - db power gain full - 46 - db cascaded noise figure full - - 11 db output ip3 full +1.5 - - dbm output p1db full -14.3 - - dbm voltage gain agc control voltage set to 20db attenuation. differential 250 ? input, differential 5k ? output load. full - 49 - db power gain full - 36 - db cascaded noise figure full - 14.1 - db output ip3 full +1.0 - - dbm output p1db full -14.4 - - dbm voltage gain agc control voltage set to 30db attenuation. differential 250 ? input, differential 5k ? output load. full - 39 - db power gain full - 26 - db cascaded noise figure full - 19.9 - db output ip3 full +0.3 - - dbm output p1db full -14.6 - - dbm voltage gain agc control voltage set to 40db attenuation. differential 250 ? input, differential 5k ? output load. full - 29 - db power gain full - 16 - db cascaded noise figure full - 27 - db output ip3 full -1.4 .74 2.8 dbm output p1db full -15.0 - - dbm voltage gain agc control voltage set to 50db attenuation. differential 250 ? input, differential 5k ? output load. full - 19 - db power gain full - 6 - db cascaded noise figure full - 35.1 - db output ip3 0-85 -2.0 - - dbm output p1db 0-85 -15.5 - - dbm voltage gain agc control voltage set to 60db attenuation. differential 250 ? input, differential 5k ? output load. full - 9 - db power gain full - -4 - db cascaded noise figure full - 43.9 - db output ip3 0-85 -3.3 - - dbm output p1db 0-85 -16.1 - - dbm voltage gain agc control voltage set to 72db attenuation. differential 250 ? input, differential 5k ? output load. full - -3 - db power gain full - -16 - db cascaded noise figure full - 60.0 - db output ip3 0-85 -6.7 - - dbm output p1db 0-85 -18.2 - - dbm minimum power gain vagc = 2.25v 25 - - -17 db agc gain control voltage full 0.2 - 2.25 v agc gain control sensitivity over supply range full - 61.6 - db/v receive cascaded ac electrical speci?ation if = 375mhz, lo = 748mhz, v cc = 2.7v, unless otherwise speci?d (continued) parameter test conditions temp. ( o c) min typ max units HFA3783
8 agc gain control input impedance full 20 23 - k ? gain switching speed to 1db settling full agc scale full - 0.4 1 s insertion phase vs agc full agc range 25 -2 0.3 +2 deg/db if detector response time 10pf, 2.9k external load full - 0.15 0.25 s if detector input voltage 0.5v, 175 a into 2.87k out full 100 150 200 mv pp lo internal input resistance single end. 748mhz 25 950 - 1.1k ? lo internal input capacitance 25 - 0.96 - pf lo drive level external 50 ? match network (single resistor) full -15 -10 0 dbm upper baseband 3db bandwidth (2nd order) full 6.7 7.4 8.5 mhz lower baseband 3db bandwidth dc coupled load full dc - - - i and q 3db bw matching full -2 - +2 % cascaded receive i or q baseband thd 1mhz, 1v pp diff. for first 50db of attenuation range 25 - - 1 % cascaded receive i/q crosstalk 25 - - -40 db i/q amplitude balance 100khz cw full -1 - +1 db i/q phase balance 100khz cw full -2 - +2 deg cascaded i or q baseband differential offset voltage after calibration cycle. measured with a setting of 26db of power gain full - - 10 mv cascaded i or q common mode voltage at baseband full 1.08 1.17 1.32 v offset calibration time ref = 44mhz, offset counter c = 25 full - 25 - s offset counter divide ratio (c counter) input ref clock is divided by c * 2 for sar offset correction full 1 - 127 - cal_en minimum pulse width high to low to high transition time full 0 - - ns baseband output resistance loading differential. 1/2 value for ground reference loads full - 5 - k ? baseband output capacitance loading single end, each full - - 10 pf differential full - - 10 pf note: 4. a positive frequency offset from the carrier produces i leading q by 90 degrees. transmit cascaded ac electrical specifications lo = 748mhz, v cc = 2.7v, vcm = 1.24v unless otherwise specified parameter test conditions temp. ( o c) min typ max units if frequency range test diagram full 70 - 600 mhz 2 x lo frequency range test diagram full 140 - 1200 mhz output power at 250 ? differential load agc voltage set to -10dbm output power for 0.35v pp sine i and q inputs full - -10 - dbm output noise floor full - -141 - dbm/hz p1db/output power ratio full 10 - - db receive cascaded ac electrical speci?ation if = 375mhz, lo = 748mhz, v cc = 2.7v, unless otherwise speci?d (continued) parameter test conditions temp. ( o c) min typ max units HFA3783
9 output power at 250 ? differential load agc voltage set to 10db attenuation. 0.35v pp sine i and q inputs full - -20 - dbm output noise floor full - -149 - dbm/hz p1db/output power ratio full 10 - - db output power at 250 ? differential load agc voltage set to 20db attenuation. 0.35v pp sine i and q inputs full - -30 - dbm output noise floor full - -157 - dbm/hz p1db/output power ratio full 10 - - db output power at 250 ? differential load agc voltage set to 30db attenuation. 0.35v pp sine i and q inputs full - -40 - dbm output noise floor full - -161 - dbm/hz p1db/output power ratio full 10 - - db output power at 250 ? differential load agc voltage set to 40db attenuation. 0.35v pp sine i and q inputs full - -50 - dbm output noise floor full - -162 - dbm/hz p1db/output power ratio full 10 - - db output power at 250 ? differential load agc voltage set to 50db attenuation. 0.35v pp sine i and q inputs full - -60 - dbm output noise floor full - -163 - dbm/hz p1db/output power ratio full 10 - - db output power at 250 ? differential load agc voltage set to 60db attenuation. 0.35v pp sine i and q inputs full - -70 - dbm output noise floor full - -164 - dbm/hz p1db/output power ratio full 10 - - db output power at 250 ? differential load agc voltage set to 70db attenuation. 0.35v pp sine i and q inputs full - -80 - dbm output noise floor full - -164 - dbm/hz p1db/output power ratio full 10 - - db agc gain control voltage full 0.1 - 2.25 v agc gain control sensitivity supply range 25 - 35.4 - db/v agc control input impedance full 20 21 - k ? gain switching speed to 1% settling full scale 25 - 0.8 4 s insertion phase vs agc 50db range from max full - - 4.0 deg i/q baseband bandwidth application circuit full 0 13 - mhz cascaded baseband to if tx thd 1mhz, 0.5v pp 25 - - 0.5 % amplitude balance dc inputs 25 -0.5 - +0.5 db phase balance dc inputs 25 -2 - +2 deg carrier suppression full agc range 25 - -43 -30 dbc ssb sideband suppression (note 5) 100khz inputs, full agc range 25 - -43 -32 dbc optimum if output differential impedance shared with rx 25 - 250 - ? lo internal input resistance single end across f. range same as rx section 25 950 - 1.1k ? lo internal input capacitance 25 - 0.96 - pf lo drive level external 50 ? match network (single resistor) full -15 -10 0 dbm baseband differential input impedance full 100 150 - k ? optimum baseband differential input voltage shaped pulses full - 0.5 - v pp common mode baseband input voltage range all tx inputs full 1.2 1.30 1.40 v note: 5. i leading q produces a+jw ccw rotation and a positive frequency offset from the carrier. transmit cascaded ac electrical specifications lo = 748mhz, v cc = 2.7v, vcm = 1.24v unless otherwise specified (continued) parameter test conditions temp. ( o c) min typ max units HFA3783
10 phase lock loop electrical speci?ations parameter test conditions temp. ( o c) min typ max units operating 2x lo frequency test diagram full 140 - 1200 mhz reference oscillator frequency test diagram full - - 50 mhz selectable prescaler ratios (2 settings) full 16/17 n/a 32/33 - swallow counter divide ratio (a counter) full 0 - 127 - programmable counter divide ratio (b counter) full 3 - 2047 - reference counter divide ratio (r counter) full 3 - 32767 - reference oscillator sensitivity single or differential sine inputs full 0.5 - - v pp cmos single or complementary full - cmos - - reference oscillator duty cycle cmos inputs full 40 - 60 % charge pump sink/source current/tolerance 250 a selection +/- 25% full 0.18 0.25 0.32 ma charge pump sink/source current/tolerance 500 a selection +/- 25% full 0.375 0.5 0.625 ma charge pump sink/source current/tolerance 750 a selection +/- 25% full 0.56 0.75 0.94 ma charge pump sink/source current/tolerance 1ma selection +/- 25% full 0.75 1.0 1.25 ma charge pump sink/source mismatch full - - 15 % charge pump output compliance full 0.5 - cpv dd - 0.5 v charge pump high z leakage high z state full -10 0.1 10 a charge pump supply voltage full 2.7 - 3.6 v serial interface clock width high level full 20 - - ns low level full 20 - - ns serial interface data/clk set-up time full 20 - - ns serial interface data/clk hold time full 10 - - ns serial interface clk/le set-up time full 20 - - ns serial interface le pulse width full 20 - - ns power enable truth table pe1 pe2 pll_pe (serial bus) status 0 0 1 power down state, pll registers in save mode, inactive pll, active serial interface 1 1 1 receive state, active pll 1 0 1 transmit state, active pll 0 1 1 inactive transmit and receive states, active pll, active serial interface x x 0 inactive pll, disabled pll registers, active serial interface pll synthesizer and dc offset clock programming table serial bits register definition lsb 1 2 345678910111213 14 15 16171819msb r counter 0 0 r(0) r(1) r(2) r(3) r(4) r(5) r(6) r(7) r(8) r(9) r(10) r(11) r(12) r(13) r(14) x (don? care) a/b counter 0 1 a(0) a(1) a(2) a(3) a(4) a(5) a(6) b(0) b(1) b(2) b(3) b(4) b(5) b(6) b(7) b(8) b(9) b(10) HFA3783
11 operational mode 1 0 m(0) 0 m(2) m(3) m(4) m(5) m(6) m(7) m(8) 0 0 0 0 m(13) m(14) m(15) x x offset calibration 1 1 c(0) c(1) c(2) c(3) c(4) c(5) c(6) 0 0 0 0 c(11) x (don? care) notes: 6. the serial data is clocked on the rising edge of the serial clock, msb first. the serial interface is active when le is low. the serial data is latched into defined registers on the rising edge of le. 7. the m register or operational mode needs to be loaded first. registers r, a/b and offset calibration follow m loading in any sequence. reference frequency counter/divider bit description r(0-14) least signi?ant bit r(0) to most signi?ant bit r(14) of the divide by r counter. the reference signal frequency is divided down by this counter and is compared with a divided lo by a phase detector. lo frequency counters/dividers bit description a(0-6) least signi?ant bit a(0) to most signi?ant bit a(6) of a 7-bit swallow counter and lsb b(0) to msb b(10) of the 11 bits divider. the lo frequency is divided down by [p * b+a], where p is the prescaler divider set by bit m(2). this divided signal frequency is compared by a phase detector with the divided reference signal. b(0-10) operational modes bit description m(0) (pll_pe), phase lock loop power enable. 1 = enable, 0 = power down. serial port always on. m(2) prescaler select. 0 = 16/17, 1 = 32/33 m(3) m(4) charge pump current setting. m(4) m(3) output sink/source 0 0 0.25ma 0 1 0.50ma 1 0 0.75ma 1 1 1.00ma m(5) m(6) charge pump sign. m(6) m(5) 0 0 source current if lo/ [p * b+a] < ref/r 0 1 source current if lo/ [p * b+a] > ref/r m(7) m(8) m(13) ld pin multiplex operation. m(13) m(8) m(7) output at pin ld 0 0 x lock detect operation 0 1 x short to gnd 1 0 x serial register read back 1 1 0 ref. divided by r waveform 1 1 1 lo divided by [p * b+a] waveform m(14) m(15) charge pump operation/test. m(15) m(14) operation/test 0 0 normal operation 0 1 charge pump constant current source 1 0 charge pump constant current sink 1 1 high impedance state pll synthesizer and dc offset clock programming table (continued) serial bits register definition lsb 1 2 345678910111213 14 15 16171819msb HFA3783
12 dc offset calibration counter bit description c(0-6) least significant bit c(0) to most significant bit c(6) of the offset calibration counter/divider. the calibration clock frequency and calibration time is defined by the reference signal frequency divided down by this counter as follows: c(11) set output bias level for ac coupling applications and tx/rx switching improvement in performance. cal time = 22 ? 2 ? c refin (mhz) ----------------------------------- - dat/clk clk width clk width clk/le le lsb bit 1 bit 2 clk data le figure 1. pll synthesizer serial interface timing diagram high set up low hold msb bit 20 p. width dat/clk set up HFA3783
13 s parameter tables rx differential input, linear mode freq (mhz) mag angle 70 0.886 -2.6 140 0.886 -4.7 200 0.886 -6.6 280 0.885 -9.4 380 0.885 -12.8 500 0.883 -16.9 600 0.883 -20.1 rx differential input, tx mode freq (mhz) mag angle 70 0.877 -4.4 140 0.873 -7.4 200 0.870 -10.5 280 0.866 -14.5 380 0.862 -19.6 500 0.857 -25.7 600 0.853 -30.5 rx differential input, saturated freq (mhz) mag angle 70 0.883 -2.5 140 0.881 -5.7 200 0.878 -8.4 280 0.875 -11.9 380 0.869 -16.2 500 0.859 -21.3 600 0.850 -25.4 tx differential output freq (mhz) mag angle 70 1 -1.1 140 1 -2.0 200 0.999 -2.8 280 0.999 -3.9 380 0.999 -5.4 500 0.999 -7.1 600 0.997 -8.3 tx diff out at rx-mode freq (mhz) mag angle 70 1 -1.0 140 1 -1.9 200 1 -2.8 280 1 -3.9 380 1 -5.2 500 0.999 -6.8 600 0.999 -8.0 lo input single end freq (mhz) mag angle 140 0.923 -5.1 400 0.920 -13.4 560 0.917 -19.0 760 0.911 -25.9 1000 0.900 -34.8 1200 0.890 -42.3 ref in single end freq (mhz) resistor /capacitance parallel 10 5.8k 0.840p 30 5.7k 0.850p 50 5.7k 0.860p rx single end in linear mode freq (mhz) mag angle 70 0.873 -4.0 140 0.872 -7.1 200 0.870 -10.1 280 0.869 -14.2 380 0.870 -19.3 500 0.872 -25.6 600 0.872 -30.8 HFA3783
14 overall device description the HFA3783 is a highly integrated baseband converter for half duplex wireless data applications. it features all the necessary blocks for baseband modulation and demodulation of ??and ??quadrature multiplexing signals including an on chip three wire interface pll stage used with an external vco for local oscillator applications. device rf properties have been optimized through the thoughtful consideration of layout, device pinout, and a completely differential design. these rf properties include immunity from common mode signals such as noise and crosstalk, optimized dynamic range for low power requirements and reduced relevant parasitics and settling times. the single power supply requirements from 2.7v dc to 3.3v dc makes the HFA3783 a good choice for portable transceiver designs. receive chain the HFA3783 has two cascaded very low distortion integrated agc if ampli?rs with frequency response from 70 to 600mhz. these differential ampli?rs exhibit better than 70db of both voltage gain and agc range. noise ?ure, output compression and intercept point variations with the agc range have been tailored to achieve cascaded performances as presented in the ac electrical speci?ations. to increase the receivers overall agc dynamic range and conserve compression speci?ations, a peak detector has been added in parallel with the agcs input. the peak detector is used to control an external step attenuator or the rf gain of the front end lna stage. following the agc stages, an ac coupled down conversion pair of quadrature doubly balanced mixers are used for ? and ??baseband if processing. these differential converters are driven by an internal differential quadrature generator with broadband response and excellent quadrature properties. for broadband operation, the local oscillator frequency input is twice the desired frequency of demodulation. duty cycle and signal purity requirements for the 2xlo input using this type of quadrature architecture are less restrictive for the HFA3783. ground reference or differential input signals from -15dbm to 0dbm and frequencies up to 1200mhz (2xlo) can be used. the output of the ? and ? mixers are dc coupled to a pair of multistage differential 2nd pole antialiasing baseband ?ters with dc offset correction. the dc offset correction is enabled with an external control pin allowing for correction to occur during transmit, receive or power down modes. the baseband ?ters cut off frequency of 7.7mhz is optimized for 11m chips/s spread spectrum applications. the baseband outputs are differential, with common mode dc voltage outputs tracking an internal band gap voltage reference. the band gap reference is also available to the user by an external pin. the ??and ??baseband voltages can swing up to 1vpp differential, following the ac electrical speci?ations across the agc range. figure 16 illustrates the cascaded gain characteristics versus agc voltage control for the HFA3783 receive section. transmit chain the HFA3783 modulator section has a frequency response of 70 to 600mhz. it consists of differential ??and ? baseband inputs requiring pre-shaped analog data levels up to 500mvpp. a common mode voltage of around 1.3v is required for proper operation of the four differential input pins. there are no internal pre-shaping ?ters in the modulator section. following the differential input stages, a dc coupled up conversion pair of quadrature doubly balanced mixers are used for ??and ??baseband if processing. these differential mixers are driven by the same internal lo quadrature generator used in the receive section. their phase and gain characteristics, including i/q matching, are well suitable for accurate data transmission. the ?al stage is an agc ampli?r with 70db of dynamic range. please refer to figure 35. detailed description receive agc/ peak detector the receive agc ampli?r section consists of 4 stages and each stage is built out of four parallel, distributed gain/degeneration differential pairs. in half duplex packet transmission linear systems, the receive agc controls thermal and supply voltage variations over the packet duration are more important than gain control linearity. therefore, the chosen architecture addresses very constricted temperature, voltage and process variations. the control is based on a band gap voltage reference ?m distribution scheme. in addition, the design provides fast agc settling times as well as fast turn on/off characteristics for packetized information. the four stage agc ampli?r has a typical maximum voltage gain of 44db and exhibits better than 70db of dynamic range, providing an attenuation in excess of 26db at minimum gain. the design can be used differential or single ended, exhibiting the same gain characteristics: however, consideration is necessary due to common mode spurious signals. one of the main features of this front end is the high impedance and small variation of s parameters when the HFA3783 is switched between transmit and receive modes. this feature permits the use of a combination match network and the use of a single saw ?ter for both halves of the duplex operation. s parameters for the differential and single ended applications are available in the s parameter tables of this document. the matching network arrangements will be discussed later in if interface section. a peak detector is placed in parallel with the input of the ?st stage of the agc ampli?r. it consists of a high frequency differential full wave recti?r and a voltage to current converter. the peak detector has limited range and is used to trip a comparator in an external baseband processor when the voltage swing at the input of the agc ampli?r is about 150mvpp. once the external comparator is tripped, its logic output level steps the lnas gain down keeping the rf HFA3783
15 and if mixers out of compression. an external resistor and capacitor set both the desired threshold voltage and time constant. figures 29 and 30 illustrate the typical current output of the peak detector for input voltage levels between 100 and 200mvpp. quadrature demodulator the output of the agc ampli?r is ac coupled to two doubly balanced quadrature differential mixers, for ??and ? demodulation. with full balanced differential architecture, these mixers are driven by an accurate internal local oscillator (lo) chain as described later. the voltage gain for both mixers is well matched with a typical value of 8v/v. low pass filter and dc offset correction to cover baseband signals from dc to 7.7mhz, the outputs of the baseband down converter mixers are dc coupled to the low pass filter stages. for true dc response, the combination of all dc offsets (mixer, lpf and buffers) needs to be calibrated for accurate baseband processing. this calibration can be performed at any time during the receive, transmit or power down modes. figure 2 depicts the baseband low pass receive ?ter implementation and figure 3 shows the calibration internal timing diagram of the HFA3783. referring to channel ??for example, calibration begins with the auto balanced comparator measuring the differential offset between the rxi+ and rxi- outputs. the comparators output is fed to a decision circuit which changes the condition of a successive approximation register (sar) state control. the sar controls 8 bits of a current output digital to analog converter (idac) which is divided by weight into a lpf section (2 pole) and a buffer ampli?r. the currents are searched and set to bring the offset to a minimum. the lpf has a xed gain of 2.5v/v and the buffer adds a 1.25v/v ?al gain to the receive chain. referring to figure 2, clocking to the sar is provided by a programmable division of the ref_in signal. (used for the pll as the stable reference.) the frequency of the reference signal is divided down by the register setting of the offset calibration counter. (details for setting this counter can be found in the programming the pll synthesizer and dc offset clock section.) the output of the calibration counter is again divided by 2 and the period used to generate the time slots of a state sequence. the calibration cycle is initialized by a rising edge on the HFA3783 cal_en pin. the state sequence slots 1 to 7 are used to settle all circuits in case the device is in the power down mode, slots 8 to 10 are used to calibrate the offset comparators (auto balancing) and slots 13 to 21 perform the search with an initial value of approximately + or - 400mv differential dc level. the comparator reads the direction and level of the offset and sets the next level and polarity at + or -400/2 mv. the process continues until slot 21 in a divide by 2 polarity and minimum offset search. the contents of the sar are kept in slot 22 which holds the idac in storage mode until a new positive edge is provided to the cal_en pin. in receive mode, the agc ampli?rs are turned off during the calibration cycle. a typical calibration time from 10 to 25 s is suggested for optimum accuracy. the baseband outputs of the lpf buffer ampli?r drive differential loads of 5k ? with a common mode voltage of typically 1.17v. an extra feature of the lpf allows for ac coupling of the baseband differential outputs. to avoid discharging of the ac coupling capacitors between transmit and receive states a common mode voltage can be applied to all outputs. an onboard programmable bit control establishes the application with 4 internal resistors and switches. lo quadrature generator the in phase and quadrature local oscillator signals are generated by a divide by two circuit that drives both the up and down conversion mixers. with a fully balanced approach, the phase relationship between the two quadrature signals is within 90 o 2 o for a wide 70 to 600mhz frequency range. the input signal frequency at the lo_in pin needs to be twice the desired local oscillator frequency. the high impedance differential lo_in+ and lo_in- inputs, which are driven by an external vco, can be used single ended by capacitively bypassing one input to ground. the user needs to terminate the vco transmission line into the desired impedance and ac couple the active lo_in input. divide by two lo generation often requires rigid control of signal purity or duty cycles. the HFA3783 has an internal duty cycle compensation circuit which eases the requirements of rigidly controlled duty cycles. second harmonic contents up to 10% are acceptable. HFA3783
16 figure 2. dc offset calibration block diagram figure 3. dc offset calibration timing diagram 8 8 idac idac sar buffer buffer lpf lpf comp comp auto bal. cal_en ref_in cal cal clk rxi+ rxi- rxq+ rxq- bits c<0:6> bit c<11> cm pin 42 counter bit c<11> pin 36 pin 35 pin 14 pin 37 pin 38 control voltage cm voltage ref/c ref/2c slot 1 slot 2 cal_en slot 8 slot 10 slot 13 1 2 3 4 5 6 7 8 slot 21 slot 22 calibrate comparators agc amp turned off in rx mode agc amp on-baseband natural agc amp on calibrated offset at baseband calibration starts at next rise time of (ref/counter) setting from the serial interface allocated settling time (cal clk) offset if cal_en is low store cal HFA3783
17 pll the HFA3783 includes a classical architecture phase lock loop circuit with a three wire serial control interface to be used with an external vco. figure 4 depicts a simpli?d block diagram of the pll. it consists of a programmable ? counter used to divide down the frequency of a very stable reference signal up to 50mhz to a phase comparator. a couple of counters (??and ?? with a front end prescaler (? or p+1?, with dual modulus control, divides down the frequency of an external vco signal to the same phase comparator. the comparator controls a charge pump circuit and an external loop ?ter closes the loop for vco control. the vco frequency dividing chain works with a dual modulus control as follows: at the beginning of a count cycle, and if the a counter is programmed with a value greater than zero, the prescaler is set to a division ratio of (p+1) where p can take programmable values of 16 or 32. notice that the prescaler output signal is always fed simultaneously to both a and b counters. upon ?ling counter a, the prescaler division ratio becomes p and the b counter continues on its own with a in standby. this process is known as ?ulse swallowing? the expression b-a (counts) is the remainder of counts carried out by the b counter after a is full. both a and b counters are reset at the end of the counting cycle when b ?ls up. as a result, the total count or division ratio used for the vco signal is a*(p+1) + (b-a)*p which simpli?s to [p*b+a]. (a and b counters are referred as the ??counter). the charge pump (current source/sink) has 4 programmable current settings. this variation allows the user to change the reference frequency for different objectives without changing the loop ?ter components. the user can program the charge pump sign based on the direction of increase or decrease of the vco frequency. the figure 4. pll simplified block diagram figure 5. charge pump output for two slightly different frequency signals a b p/p+1 n counter prescaler reset reset dual modulus r r counter to lo divide by 2 drivers vco vcontrol isource isink v ref_in cp_d0 lo_in+ to pin 14 control pin 27 or 26 pin 22 dc offset cal vco [p*b+a] ref r 1/2v cc HFA3783
18 most often used vcos in the market have positive kvcos where the vco frequency increases with an increase in control voltage. in this case, the charge pump current shall ?ource?current (to the main capacitor of the loop ?ter) when the vco frequency becomes less than the desired frequency of operation. the phase comparison and charge pump output behavior in a open loop system is illustrated in figure 5. the comparators inputs (the top two waveforms of figure 5 are from the n and r counters. the output from the ??counter and the prescaler, labelled as ?co/[p*b+a] shows a lower frequency than the output from the ? counter labeled ?ef/r? ref/r is usually called ?eference frequency. the bottom waveform represents the charge pump sourcing current as it has been programmed. because it is an open loop system, the charge pump current pulse width will increase and follow the phase comparators output. the charge pump signal can be developed across a resistor connected between pin 22 and a power supply of half the v cc voltage. in the case where the vco/[p*b+a] frequency is higher than the ref/r frequency, the bottom waveform would have negative pulse width variations indicating the charge pump sinking current. the closed loop concept can be understood intuitively by observing the bottom waveform and noticing the tendency of the charge pump to ?harge?a capacitor (loop ?ter) and increase the vco voltage control accordingly. as the vco/[p*b+a] frequency becomes higher than the ref/r frequency, the charge pump begins to sink current and the vco control voltage begins to drop. the process would continue in equilibrium with expected sharp reverting polarity pulses at the ref/r reference frequency. figure 6 depicts a simple charge pump polarity concept and includes the output of the lock detect pin of the HFA3783. this pin has other applications and will be covered in the next section. pll synthesizer and dc offset clock programming a three wire cmos serial interface (clk, data, le) programs various counters and operational modes of the HFA3783 pll. it also programs the dc offset adjust counter and operation of the lpf section. figure 1 in the speci?ation section shows the timing diagram for this interface. short clock periods in the order of 20ns can be used to program this interface. the serial data is clocked on the rising edge of the serial clock into a serial 20-bit shift register with the msb ?st. see the pll synthesizer and dc clock programming table for details. the serial register is always active when the le pin is held low. on the rising edge of the le pin, the serial register is loaded and latched into the addressed registers for the particular function. the two least signi?ant bits address the intended register for loading the serial data. this interface has been designed for a minimum le pulse width. there is no need to discontinue the clock during loading of the 4 intended registers. note: upon a rising edge on le, the HFA3783 pll unlocks the loop during a random period varying from 0 to 1/(reference frequency). fast frequency hopping applications may be affected during this time. ref n cp ld figure 6. simplified cp and lock detect output waveforms HFA3783
19 the four registers are as follows: r counter: division factor ??in binary weight format with r(0) as 2 0 and so on, for a decimal integer division ratio for the stable reference signal. a/b counter: a combination of binary weighted integer division factors for the ??counter as explained by the relationship p*b+a. operational mode: these register bits control the charge pump operation, prescaler ??setting, the power down feature of the pll and the functions of the ld output pin. offset calibration: these register bits control the division ratio, in binary weight, for the sar clock and a special baseband output state for the low pass filter. note: at power up (v cc application), it is important to load the operational mode register before any sequence of the remaining registers. operational modes description bit m(0): this bit is normally set at one for the pll operation. setting to zero can save up to 6ma of supply current by disabling the pll, although the serial interface is always active for loading data. this operational mode bit controls the serial interface at power up and it is important to be loaded ?st, after application of v cc . bit m(2): selects the prescaler ??for either 16 or 32. bits m(3),m(4): these bits select the desired charge pump current from 250 a to 1ma in four steps. bits m(5), m(6): programming 00 will set the charge pump to ?ource?current when the vco frequency is below the desired frequency. it is used for vcos where the frequency increases with increase in the voltage control. programming 01 sets the charge pump to sink current when the vco frequency is below the desired frequency. it is used for vcos where the frequency increases with decrease in the voltage control (negative kvco). bits m(8), m(7) and m(13): these bits de?e the ld output multiple operation. during the lock detect operation, the ld output follows the phase comparator output and can be used with external integration, as a frequency lock monitor function. ld output can be shorted to ground or used as a monitor pin for either the output of the ??counter divider or the [p*b+a] dual modulus divider. in addition, it can be used as the serial register read back for testing purposes in a fifo mode (not the latched register/counters themselves) by reading the msb on the falling edge of le and the remaining bits on the rising clk edges. bits m(14), m(15): these bits set the charge pump operation for normal operation, constant sink or source and in a high impedance state. the high impedance state allows for external control. dc offset calibration counter description bits c(0) to c(6): set a binary weighted decimal integer number for the stable reference input frequency division ratio. the ratio is used by the sar for dc offset calibration in the HFA3783 and previously described in the low pass filters section of this document. bit c(11): enables a dc hold circuit which allows ac coupling of the baseband signals to a processor a/ds. a common mode voltage applied to the baseband outputs during transmit mode switching reduces the coupling capacitors charging times. quadrature modulator the differential baseband signals for the HFA3783 modulator require a controlled common mode voltage for proper operation of the device. carrier suppression is consequently a function of the common mode dc match between the differential legs of each of the ??and ? channels. the modulator bandwidth is very wide and need to be limited by external means. the inputs are equivalent to driving the up conversion quadrature mixers directly; therefore provisions for shaping the baseband signals before up conversion have to be made externally. shaping can be accomplished either by an external ?ter or by pre-shaping in a baseband processor. baseband signals up to 500mvpp differential can be used at the ??and ??ports. centered upon a common mode voltage, the 500mvpp pre- shaped differential signals were used for the compression characteristics speci?d in this document. by reducing the magnitude of these signals improved low distortion modulation characteristics can be realized. the quiescent current for the upconversion mixers is established by the common mode input dc signal. by setting the common mode voltage to zero during the receive mode, power dissipation and mixer noise in the transmit path is reduced. the common mode voltage, routed through the baseband processor for temperature and v cc tracking, is normally established by the HFA3783s on board 1.2v reference. this reference is inactive during the power down mode. the quadrature up converter mixers are also of a doubly balanced design. ??and ??up converter signals are summed and buffered to drive the next stage, the agc ampli?r. as with the demodulators, both modulator mixers are driven from the same quadrature lo generator. these mixers feature a phase balance of 2 o and amplitude balance of 0.5db from 70 to 600mhz. these qualities are re?cted into the ssb characteristics. for differential ??and ?? 100khz sinusoidal inputs of 375mvpp, 90 o apart, the carrier feedthrough is typical -43dbc with typical sideband suppression of 43dbc at 374mhz. a differential open collector linear output agc ampli?r with 70db of dynamic range follows the mixers. this ampli?r is based in a tight controlled voltage and temperature current HFA3783
20 steering mechanism for gain control. the ampli?r main function is controlling the power output of the transmit signal and has very linear agc characteristics as shown in figure 35. the differential open collector outputs require v cc biasing as with any open collector application and exhibit high isolation. the HFA3783 output impedance is constant whether in the receive or transmit mode. consequently, a combination matching network with the use of a single saw ?ter can be used for both halves of the duplex operation. single ended operation is discouraged due to; tx and rx return loss variation, loss of power output and lack of cancellation of pll induced spurious signals. differential summing match networks are strongly recommended when using single end saw devices. s parameters for the output port are available in the s parameter tables section. the agc ampli?r feature an output compression level of 1v p-p , with a cascaded performance capable of generating a typical cw power of -10dbm into 250 ? when differential inputs of 250mv dc are applied to both ??and ??inputs. if interface both modulator and demodulator of the HFA3783 ac cascaded speci?ations in this document were characterized in a 250 ? system. the high impedance of the receive input and the open collector output structure of the transmit channel permit the use of a combination match network capable of interfacing with only one differential ?ter device in duplex operation. in addition, the HFA3783 input and output impedances have small variations when the device changes its mode of operation from transmit to receive. the system impedance (250 ? ) is de?ed by the ?ter input/output impedance including its own match networks and this value has been chosen as a compromise between current consumption, voltage swing and therefore compression. a higher system zo can compromise the voltage swing capabilities due to the low voltage operation of the HFA3783 and a low system zo affects the power supply current consumed by the application in general, for the same rf power budget. the output match network of the transmit output, includes a differential ??match network used to bias the differential collectors which are of high impedance. this high impedance is lowered to a value of around 2k ? by a parallel resistor placed across the collector terminals. this value sets the output impedance of the two collectors and also serves as a compromise value for the loaded ??of the network for a desired system bandwidth. the other side of the match network is set to match 250 ? (from a ?ter match application) and is directly connected to the receive differential terminals; therefore presenting a controlled termination to the high input impedance port of the receive agc. the use of dc blocking capacitors is needed to avoid a dc path between the HFA3783 receive terminals and is maybe optional depending of the differential network used to match an external ?ter to a 250 ? system. as with any differential network, symmetry is paramount. the use of matched length lines and good differential isolation, helps the structure reject common mode induced signals from other parts of the system. special attention to the collector outputs is necessary to reject v cc induced spurious signals and to reject internally induced pll spurious tones. although the network topology is simple theoretically, its implementation is challenged by layout routing and parasitics which have to be taken into consideration. ? filter match filter 250 ? 250 ? avoid ground return v cc pin 3 pin 4 pin 8 pin 9 HFA3783 figure 7. simplified if input/output combined match network network for v cc bypass close to pin 5 gnd. ? HFA3783
21 typical performance curves figure 8. rx icc vs power gain over temperature figure 9. tx icc with txi/q = 1.3v over temperature and voltage figure 10. standby icc vs v cc figure 11. 1.2v v ref voltage over v cc and temperature figure 12. charge pump 250 a setting sink and source current over temperature and voltage figure 13. charge pump 1ma setting sink and source current over temperature and voltage 39 38 37 36 35 34 33 32 31 30 29 28 27 26 25 24 23 -15 -20 -10 -5 0 5 10 15 20 25 30 35 40 45 50 55 60 +85, 3.3v +25, 3.3v +85, 2.7v +25, 2.7v -40, 3.3v -40, 2.7v rx icc (ma) rx power gain (db) 33 32 31 30 29 28 27 26 25 24 -40 20 30 80 90 3.3v 2.7v temperature (c) tx icc (ma) +85 +25 -40 160 140 120 100 80 60 40 20 0 standby icc ( a) 2.7 2.8 2.9 3.0 3.1 3.2 3.3 v cc 1.1990 1.1980 1.1970 1.1960 1.1950 1.1940 1.1930 1.1920 -40 20 30 80 90 temperature (c) v ref (v) 3.3v 2.7v 244 242 240 238 236 234 232 230 228 226 224 222 220 250 a setting ( a) -40 20 30 80 90 temperature ( o c) 2.7v, source 2.7v, sink 3.6v, sink 3.6v, source 0.99 0.97 0.95 0.93 0.91 0.89 0.87 0.85 -40 20 30 80 90 temperature ( o c) 3.6v sink 2.7v sink 2.7v source 3.6v, source 1ma setting (ma) HFA3783
22 figure 14. charge pump characteristics at 250 a figure 15. charge pump characteristics at 1ma figure 16. rx agc power gain vs vagc over temperature at all v cc typical performance curves (continued) 0.3 0.2 0.1 0 -0.1 -0.2 -0.3 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 cp current cp voltage 2.7v 3.3v 2.7v 3.3v 1.5 1.0 0.5 0 -0.5 -1.0 -1.5 1.5 1.0 0.5 0 2.0 2.5 3.0 3.5 cp voltage cp current 3.3v 2.7v 2.7v 3.3v -20 65 60 55 50 45 40 35 30 15 25 20 10 5 0 -5 -10 -15 0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 rx gain (db) vagc (v) +85 +25 -40 HFA3783
23 figure 17. rx baseband lpf profile figure 18. rx baseband spectrum, tone at 1.5mhz power gain of 56db. output converted to single ended 50 ? figure 19. rx baseband spectrum, tone at 1.5mhz power gain of -16db. output converted to single ended 50 ? typical performance curves (continued) amplitude delay 0.0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 relative scale amp, 1db/div delay, 10ns/div rbw, 300hz frequency (mhz) 10db/div 10khz 15mhz frequency ref 4.0dbm res bw = 100khz video bw = 1khz 10db/div ref 4.0dbm res bw = 100khz 15mhz 10khz frequency video bw = 1khz HFA3783
24 figure 20. rx i/q channel gain match vs power over temperature and v cc figure 21. rx i, q channel phase match vs power gain over temperature and v cc typical performance curves (continued) -20-10 0 10203040506070 rx power gain gain match variation (db) +85, 3.3v +85, 2.7v +25, 2.7v, 3.3v -40, 2.7v -40, 3.3v 0.01db/div -40, 2.7v +25, 2.7v -40, 2.7v +25, 3.3v +85, 3.3v +85, 2.7v -20 -10 0 10 20 30 40 50 60 70 rx power gain (db) phase match variation (deg) 0.05 deg/div HFA3783
25 figure 22. rx insertion phase vs vagc figure 23. rx baseband agc response time, 0dbm input figure 24. rx baseband agc response time, 0dbm input figure 25. tx to rx baseband switching time figure 26. rx to tx baseband switching time figure 27. rx baseband at power up typical performance curves (continued) 0 10 20 30 40 50 60 70 80 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 degrees (relative) vagc (v) vagc ch1 1.00v ch2 1.00v 100ns/div bb (nominal) bb (nominal) vagc ch2 1.00v 100ns/div ch1, 1.00v ch1 pe1 = 1 ch2, 2.00v 100ns/div ch1, 500mv bb (nominal) ch2 (pe2) ch1, 500mv ch2, 2.00v 100ns/div bb (nominal) pe2 pe1 = 1 ch1 ch2 ch1 ch2, pe1 pe2 = 1 ch2, 2.00v 100ns/div ch1, 500mv bb (nominal) HFA3783
26 figure 28. rx baseband at power down figure 29. if detector output current, 3 sigma distribution at all temperature and v cc figure 30. typical if detector output current at all v cc figure 31. if detector response, rise time figure 32. if detector response, fall time typical performance curves (continued) ch1 ch2 2.00v 100ns/div ch1, 500mv bb (nominal) pe2 = 0 ch2, pe1 100 120 140 160 180 200 -3 +3 300 250 200 150 100 50 0 output current input level at 374mhz, (mv pp ) distribution ( a) 250 200 150 100 50 0 100 120 140 160 180 200 input signal at 374mhz, (mv pp ) +85 output current ( a) -40 +25 if input (374mhz) 50mv/div if det output 200mv/div 50ns/div 50ns/div if det output 200mv/div 50mv/div if input (374mhz) HFA3783
27 figure 33. baseband output offset voltage variation vs vagc, if = 0v figure 34. cascaded rx frequency response, bb at 1mhz typical performance curves (continued) 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 0.5mv/div calibrated offset variation vagc (v) 0 -4 -6 70 170 270 370 470 570 670 770 870 -2 -8 relative bb output (db) frequency (mhz) HFA3783
28 figure 35. tx power out vs tx vagc over temperature at all v cc figure 36. tx ssb output characteristics at full gain figure 37. tx ssb output characteristics at full gain and wide spectrum with match network typical performance curves (continued) -5 -10 -15 -20 -25 -30 -35 -40 -45 -50 -55 -60 -65 -70 -75 -80 -85 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 -90 +85 +25 -40 tx vagc (v) ssb tx output power (dbm) 10db/div center freq = 374mhz span = 1mhz res bw = 3.0khz vbw = 3.0khz ref -8.0dbm ref -8.0dbm 10db/div 375mhz (ssb) start freq = 0.005ghz stop freq = 2.55ghz res bw = 100khz vbw = 30khz HFA3783
29 figure 38. tx ssb output characteristics at -60db from full gain figure 39. tx spread spectrum output characteristics at full gain, bb inputs at 500mv pp figure 40. tx spread spectrum output characteristics at -70db from full gain, bb inputs at 500mv pp typical performance curves (continued) 10db/div center freq = 374mhz span = 1mhz res bw = 3khz vbw = 3khz preamp gain = 50db ref -68.0dbm ref -15.0dbm 10db/div cntr freq = 374mhz span = 50mhz res bw = 3khz vbw = 100khz center freq = 374mhz span = 50mhz res bw = 300khz vbw = 100khz preamp gain = 50db ref -85.0dbm 10db/div HFA3783
30 figure 41. typical tx carrier suppression vs vagc over temperature figure 42. typical tx lower side band suppression vs vagc over temperature typical performance curves (continued) carrier suppression (dbc) -47.0 -46.5 -46.0 -45.5 -45.0 -44.5 -44.0 -43.5 -43.0 -42.5 -42.0 -41.5 -41.0 -40.5 0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 vagc (v) +25, 2.7v and 3.3v -40, 3.3v +85, 3.3v +85, 2.7v -40, 2.7v -40 -41 -42 -43 -44 -45 -46 -47 -48 -49 0.2 0 0.4 sideband suppression (dbc) 0.6 0.8 1.0 1.2 1.4 1.6 vagc (v) +85, 2.7v +85, 3.3v +25, 2.7v +25, 3.3v -40, 3.3v -40, 2.7v HFA3783
31 figure 43. typical tx carrier static amplitude and phase balance at 250mv dc differential bb inputs figure 44. tx insertion phase vs vagc figure 45. tx agc response time, full gain figure 46. tx agc response time, full gain figure 47. rx to tx if output switching time figure 48. tx to rx if output switching time typical performance curves (continued) 0.25 0.20 0.15 0.10 -0.05 -0.10 -0.15 -0.20 -0.25 -150 -0.8 -0.6 -0.4 -0.2 0 0.2 0.4 0.6 0.8 0.05 0 -100 -50 0 50 100 150 200 amp error nominal angle amp error (db) phase error (deg) phase error 90 80 70 60 50 40 30 20 10 0 -10 0 0.5 1.0 1.5 2.0 2.5 3.0 vagc (v) insertion phase, deg (relative) ch2 vagc ch1 ch1, 200mv ch2, 1.00v 200ns/div if output ch1, 200mv ch2, 1.00v 200ns/div vagc ch2 ch1 if output if output at full gain ch1, 200mv ch2 2.00v 50ns/div ch1 ch2 pe2 pe1 = 1 if output at full gain pe2 pe1 = 1 ch1 ch2 ch1, 200mv ch2 2.00v 50ns/div HFA3783
32 figure 49. tx if output at power up figure 50. tx if output at power down figure 51. tx out power vs frequency, bb at dc figure 52. eval board typical synthesizer close in phase noise figure 53. eval board typical synthesizer output with pll at 10khz bw figure 54. eval board typical synthesizer output with pll at 1khz bw typical performance curves (continued) if output at full gain ch1 ch2 pe2 = 0 pe1 ch1, 200mv ch2 2.00v 50ns/div ch1 ch2 pe1 pe2 = 0 ch1 200mv ch2 2.00v 50.0ns/div carrier power (dbm) -12 -13 -14 -15 -16 -17 -18 -19 70 170 270 370 470 570 670 770 870 frequency (mhz) refer to test diagram -30 -40 -50 -60 -70 -80 -90 -100 -110 -120 -130 ctr freq = 748khz span = 5khz res bw = 100hz vbw = 100hz -75.5dbc/hz ref level -30dbm ctr freq = 748mhz span = 100khz res bw = 1khz vbw = 100hz -30 -40 -50 -60 -70 -80 -90 -100 -110 -120 -130 ref level -30dbm ctr freq = 748mhz span = 100khz res bw = 1khz vbw = 100hz ref level -30dbm -30 -40 -50 -60 -70 -80 -90 -100 -110 -120 -130 HFA3783
33 figure 55. eval board synthesizer tx to rx switching spurious response at 1khz switching frequency, pll bw = 10khz typical performance curves (continued) -30 -40 -50 -60 -70 -80 -90 -100 -110 -120 -130 ctr freq = 748khz span = 10khz res bw = 100hz vbw = 10hz ref level -30dbm HFA3783
34 all intersil semiconductor products are manufactured, assembled and tested under iso9000 quality systems certi?ation. intersil semiconductor products are sold by description only. intersil corporation reserves the right to make changes in circuit design and/or spec ifications at any time with- out notice. accordingly, the reader is cautioned to verify that data sheets are current before placing orders. information furnished by intersil is b elieved to be accurate and reliable. however, no responsibility is assumed by intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of th ird parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of intersil or its subsidiari es. for information regarding intersil corporation and its products, see web site www.intersil.com sales of?e headquarters north america intersil corporation p. o. box 883, mail stop 53-204 melbourne, fl 32902 tel: (321) 724-7000 fax: (321) 724-7240 europe intersil sa mercure center 100, rue de la fusee 1130 brussels, belgium tel: (32) 2.724.2111 fax: (32) 2.724.22.05 asia intersil (taiwan) ltd. 7f-6, no. 101 fu hsing north road taipei, taiwan republic of china tel: (886) 2 2716 9310 fax: (886) 2 2715 3029 HFA3783 thin plastic quad flatpack packages (lqfp) d d1 e e1 -a- pin 1 a2 a1 a 11 o -13 o 11 o -13 o 0 o -7 o 0.020 0.008 min l 0 o min plane b 0.004/0.008 0.09/0.20 with plating base metal seating 0.004/0.006 0.09/0.16 b1 -b- e 0.003 0.08 a-b s d s c m 0.08 0.003 -c- -d- -h- 0.25 0.010 gage plane q48.7x7a (jedec ms-026bbc issue b) 48 lead thin plastic quad flatpack package symbol inches millimeters notes min max min max a - 0.062 - 1.60 - a1 0.002 0.005 0.05 0.15 - a2 0.054 0.057 1.35 1.45 - b 0.007 0.010 0.17 0.27 6 b1 0.007 0.009 0.17 0.23 - d 0.350 0.358 8.90 9.10 3 d1 0.272 0.280 6.90 7.10 4, 5 e 0.350 0.358 8.90 9.10 3 e1 0.272 0.280 6.90 7.10 4, 5 l 0.018 0.029 0.45 0.75 - n48 487 e 0.020 bsc 0.50 bsc - rev. 2 1/99 notes: 1. controlling dimension: millimeter. converted inch dimensions are not necessarily exact. 2. all dimensions and tolerances per ansi y14.5m-1982. 3. dimensions d and e to be determined at seating plane . 4. dimensions d1 and e1 to be determined at datum plane . 5. dimensions d1 and e1 do not include mold protrusion. allowable protrusion is 0.25mm (0.010 inch) per side. 6. dimension b does not include dambar protrusion. allowable dambar protrusion shall not cause the lead width to exceed the maximum b dimension by more than 0.08mm (0.003 inch). 7. ??is the number of terminal positions. -c- -h-

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